|Lowering distortion in power circuits without compromising their transient
response remains a primary problem for designers of audio power amplifiers. Until fairly
recently, the favorite technique for removing distortion components in linear amplifiers
was to cascade many gain stages to form a circuit having enormous amounts of gain and then
using negative feedback to control the system and correct for the many errors introduced
by this large number of components.
While the sum of these components' distortions may
cause large complex nonlinearities, the correspondingly large amounts of feedback applied
are generally more than equal to the task of cleaning up the performance with only one
trade-offthe high frequency performance of the system. Because each amplifying
device also contributes its own high frequency roll-off, and because the sum of many of
these roll-offs creates a complex, multi-pole phase lag, a system using large amounts of
negative feedback tends to be unstable at high frequencies, resulting in phenomena
popularly referred to as Transient Intermodulation Distortion (TIM). As this phenomena has
been well described elsewhere, it will be sufficient here to point out that two solutions
to TIM problems exist. The first solution is to not require any high frequency performance
of the circuit, that is, not to feed it high frequency signals it cannot handle. While
this solution works very well for many operational amplifier applications requiring only
low frequency performance, it is judged to be unacceptable in high-fidelity applications
where frequency response is required beyond 100 kiloHertz. Although human hearing is
generally very poor above 20,000 Hertz, ultrasonic frequency roll-offs produce phase and
amplitude effects in the audible region; for example, a single pole (6dB/octave) roll-off
at 30 kHz produces about 9 phase lag and 0.5 dB loss at 10 kHz. The effects may be subtle,
but their audibility is undesirable in a piece of equipment whose performance is judged by
Because of this bandwidth requirement, designers of state-of-the-art amplifiers are
turning to the other solution; simple circuits having few amplifying devices and
relatively low open loop gain. The simplicity and low gain allows the circuitry to respond
to signals very quickly, thus eliminating transient problems, but it does so at the
expense of higher harmonic and intermodulation distortions.
Because these distortions are more "musical" (having low orders of harmonics
and intermodulation sidebands), they are less offensive than TIM effects, whose high order
sidebands bear less resemblance to the naturally occurring harmonics in the music. Musical
or not, the lower order harmonics and sidebands still deserve to be removed, and the
attention of the best designers has turned to removing the distortions in the individual
amplifying devices themselves, instead of applying corrective feedback to the system.
To understand the approach to this problem, it is first necessary to note that all
distortions arise when the gain of an amplifying device is altered. A perfectly linear
device has a transfer curve which is a perfectly straight line. Any deviations
(distortion) from this straight line is the result of a gain factor which varies depending
upon the operating conditions. In real life, the gain of a transistor, tube, or FET
changes as the voltage across the device changes and as the current through the device
changes. As these conditions fluctuate, the device generates distortion, but if we hold
these conditions to a constant, the device becomes distortionless. Figure 1 is a characteristic curve of an
ideal distortionless transistor, showing absolute linearity under all conditions, whereas,
Fig. 2 is the characteristic curve of
an actual transistor. Notice that the spacing between the parallel lines is unequal,
reflecting gain changes with different currents through the transistor, and that they are
curved off the horizontal axis, showing gain changes dependent on the voltage across the
device. As the transistor wanders through these regions in reproducing the audio signal,
its gain alters, causing both harmonic and intermodulation distortion effects. If we can
limit the region of operation on this curve, particularly to the area away from the
boundaries, the distortion will be significantly reduced.
Recently, the most effective method employed for reducing distortion without feedback
has been the use of class-A operation, in which the amplifying devices are idled at very
high currents, keeping the transistor in a region on the curve where the nonlinearities
are less spectacular, as shown in Fig. 3.
While the characteristics of the transistor are less than perfect, the distortions within
the boundaries shown are relatively mild as compared with the more abrupt gain changes
outside of the dotted lines.
At great expense of efficiency, class-A operation reduces nonlinearities due to current
fluctuations through the transistor. However, it does not affect nonlinearities in the
transistor due to voltage changes. There is a method for eliminating such nonlinearities
called cascode operation, where the voltage across the transistor, tubes, or FETS is
frozen at a constant value, completely eliminating voltage-induced distortions. In the
case of transistors, the gain device can be operated in common-emitter or common-collector
modes that utilizes a second transistor in the common base mode whose emitter is connected
to the collector of the gain transistor, as in Fig.
4. Having essentially unity current gain, extremely wide bandwidth, and no
distortion, the common base device shields the gain transistor from voltage changes in the
circuit. Figure 5 shows the operating
boundaries of such a system, where the operating voltage is frozen to a constant. Figure 6 shows the effective transfer
characteristics of such a system, and we see that it more nearly approximates the curves
of the ideal transistor in Fig. 1.
A graphic demonstration of the effectiveness of such an arrangement is clearly
illustrated by the spectral analysis of a class-A emitter-follower operated without
feedback. The circuits in Fig. 7 a & b
were operated at 15 kHz at +/-5 volts. The spectral analysis of the outputs of each
circuit are shown in Fig. 8 a, b, & c,
where the vertical scale is 10 dB per division (80 dB), the horizontal scale is 0-100 kHz
at 10 kHz division, and as can be easily seen, the cascode operation of the same
transistor under otherwise identical conditions results in the reduction of distortion
from several per cent to the residual of the test setup.
Besides eliminating voltage caused nonlinearities, cascode operation can yield an
additional benefit in increased bandwidth. Because the collector-base voltage is held
constant there is minimal charging of the collector-base junction capacitance in the
transistor. Eliminating the effects of this internal lag capacitance allows higher
frequency response, thus cascode circuitry is commonly found in ultra-high frequency
amplifiers and wide bandwidth oscilloscopes where response is required beyond 100
megaHertz. Cascode circuitry has also found its way into preamplifier circuitry as
manufactured by Dayton-Wright Paragon, DB Systems, and Audio Directions among others.
With all these factors in mind, and noting that the output transistors in power
amplifiers would enjoy the beneficial effects of cascode operation, we recently undertook
the design of a cascode audio power amplifier (Patent pending) where the gain stages and
emitter-follower output stages are operated at constant voltages. The conceptual schematic
of such a device can be seen from Fig. 9,
which serves to illustrate the use of cascode operation on both the common-emitter voltage
gain stage and the common-collector output stage. In this circuit, Q1 is the input
transistor, held at a constant voltage by Q2. Q3 and Q4 form the cascode common-emitter,
voltage-gain stage which generates the full voltage swing of the amplifier. Both parts of
the circuit are biased using constant current sources, I1, I2 seen near the negative
supply rail. Output current gain is supplied by the complementary common collector
darlingtons formed by Q5-8, and Q9 and Q10 are the common base transistors which hold them
at constant voltages. V1-5 are constant voltage sources ranging from two to 10 volts. The
voltage sources on the cascode circuits can be generated by a number of arbitrary means,
including zener diodes, resistors, or even batteries.
Because voltage-induced nonlinearities take the form of "compressive"
intermodulation, it was not surprising to discover the sonic effects of utilizing cascode
operation throughout a power amplifying system corresponded to an impression of a dynamic
range capability considerably beyond what the rated power would suggest. This effect is
pronounced at high transient levels and imparts a sense of effortlessness in the
reproduction of demanding material.
While the distortion characteristics of a fully cascode amplifier are not equivalent to
those obtained through class-A operation, the lack of signal compression produces a
subjective '`ease" to the reproduced sound that closely approximates that of the
smooth nonlinearities which characterize class-A operation and are achieved without the
cost penalties attendant to a class-A output stage.